A CMOS 3.3–8.4 GHz Wide Tuning Range, Low Phase Noise LC VCO Bodhisatwa Sadhu, Jaehyup Kim and Ramesh Harjani University of Minnesota, Minneapolis, MN 55455, Email: harjani@ece.umn.edu Abstract—A novel inductor switching technique is used to de- sign and implement a wideband LC voltage controlled oscillator (VCO) in 0.13μm CMOS. The VCO has a tuning range of 87.2% between 3.3 and 8.4 GHz with phase noise ranging from -122 to -117.2 dBc/Hz at 1MHz offset. The power varies between 6.5 and 15.4 mW over the tuning range. This results in a Power- Frequency-Tuning Normalized figure of merit (PFTN) between 6.6 and 10.2 dB which is one of the best reported to date. I. I NTRODUCTION With increasing congestion in the radio spectrum, multi- band multi-standard transceivers have gained prominence. These broadband systems require VCOs that achieve a wide tuning range coupled with stringent phase noise performance. Although LC VCOs are traditionally well-suited for low phase noise applications, their tuning range remains notoriously limited. For LC VCOs that employ only capacitance switching and/or tuning, the lowest achievable frequency is limited by the start-up criteria, and the highest achievable frequency is limited by the parasitic capacitance of the oscillator. Addi- tionally, these systems are plagued by increased power at low frequencies and increased phase noise at high frequencies. As will be discussed in this paper, switching the inductance in addition to varying the capacitance eliminates these problems and provides a viable solution for obtaining very wide tuning ranges alongside low phase noise performance. The paper is organized as follows. We discuss the advan- tages of inductance switching in Section II and describe a prototype design in Section III. We enumerate and discuss the experimental results in Section IV, and finally draw our conclusions in Section V. II. TUNING OPTIONS IN LC TANK VCOS Frequency variation in an LC tank can be realized by changing the capacitance and/or the inductance of the tank. Owing to a difficulty in the physical realization of high performance tuned inductors, pure inductive tuning is almost unknown. Capacitive tuning using varactors, or a combination of switched capacitors and varactors has been used frequently to obtain a wide tuning range. However, for a very wide-tuning range, pure capacitive tuning is not optimal. Instead the in- ductance may be switched [1]–[3] in order to provide separate frequency banks. Within these banks, frequency tuning may be accomplished through capacitive tuning. A detailed description regarding the tradeoffs associated with designing VCOs with inductor bands has be provided in [3] and is not provided here for space limitations. A combined switching and tuning scheme, as described above, provides significant advantages over pure capacitive Frequency (log scale) Power (log scale) Constant voltage swing regime Start-up limited regime Worst case power dissipation Frequency (log scale) Power (log scale) Worst case power dissipation L1 L2 L3 L4 L1 > L2 > L3 > L4 (a) Tank A (b) Tank B Fig. 1. A comparison of power consumption between (a) tank A: an unswitched inductor resonator, and (b) tank B: a switched inductor resonator tuning for very wide tuning ranges. The reason can be ex- plained through a simple comparison. Consider two different LC tanks: tank A where the frequency is tuned exclusively using capacitance variation, and tank B where the inductance is switched in addition to capacitive tuning. For a constant voltage swing, the power dissipation of an LC tank VCO is given by equation (1), where R s is the parasitic series resistance of the inductor and V sw is the voltage swing. V sw = (ωL) 2 R s I bias = (ωL) 2 R s P ower V DD ⇒ P ower = R s V sw V DD (ωL) 2 (1) Fig. 1 shows a plot of the power dissipation versus frequency for tanks A and B. As expressed in (1), the power dissi- pated for a constant voltage swing decreases with increasing frequency (slope of -2 on a log-log plot). Moreover, for tank A, Barkhausen’s criteria determines the current at the lower frequencies causing a steeper decrease in power with increasing frequency (slope of -4 on a log-log plot). However, in the case of tank B, since the power dissipated is inversely proportional to the inductance as expressed in (1), using a larger value of inductance in the lower frequency ranges causes a significant drop in power. Comparing the two tanks, the worst case power dissipation is significantly minimized in tank B as shown in Fig 1. A similar observation can be drawn in the case of the phase noise of the VCO. From Leeson’s model, the approximate phase noise of the VCO is given by (2), which for ω 0 ≫ QΔω, reduces to (3). Here Δω is the frequency offset for the phase noise measurement, F is the noise factor, R s is the parasitic series resistance of the inductor, P sig is the signal power, ω 0 is the oscillation frequency, and Q is the inductor’s 559 IEEE 2009 Custom Intergrated Circuits Conference (CICC) 978-1-4244-4072-6/09/$25.00 ©2009 IEEE T-24-1